60 GHz Wideband Class E/F2 Power Amplifier

ABSTRACT

A novel and useful fully integrated switched-mode wideband 60 GHz power amplifier architecture. Using an appropriate second-harmonic termination of its output matching network, the required systematic peak current of the final stage is reduced such that the PA functions as a class-E/F 2  switched-mode PA at saturation. In addition, low/moderate magnetic coupling factor transformers in the intermediate stages enable the PA to reach a high power added efficiency (PAE) and bandwidth product. Transformers of low/moderate coupling are also utilized in the preliminary stages of the PA to improve the overall bandwidth. In addition, the PA exploits the different behavior of the output impedance matching network for differential mode (DM) and common mode (CM) excitations. The PA is also stabilized against the combination of DM and CM oscillation modes. The PA also provides a technique to stabilize transformer-based mm-wave amplifiers against various modes of undesired oscillations.

REFERENCE TO PRIORITY APPLICATION

This application claims priority to U.S. Provisional Application Ser.No. 62/059,448, filed Oct. 3, 2014, entitled “Power Amplifier,”incorporated herein by reference in its entirety.

FIELD OF THE INVENTION

The present invention relates generally to amplifier circuits and inparticular to a 60 GHz wideband class E/F₂ power amplifier.

BACKGROUND OF THE INVENTION

Radio frequency (RF) power amplifier is a key component of radios thatfunctions to convert a low power RF signal into a larger signal havingsignificant power for driving, for example, the antenna of atransmitter. As more and more of the radio processing chain goesdigital, there is currently great effort to digitize and integrate thePA, which traditionally was implemented off-chip with discrete analogcomponents.

Considering digital radios, a key technical challenge of 60 GHz CMOSradios is poor efficiency of their power amplifier (PA). By usingadvanced digitally intensive transmitter architectures, such asoutphasing and direct digital-to-RF conversion, a nonlinear switch-modePA can be used to improve the total system efficiency. Switching poweramplifiers, however, are rarely seen at mm-wave frequencies due to thelarge output capacitance and low current capability of CMOS transistors.In a prior art power amplifier, a two-turn inductor is exploited torealize a switch-mode PA at 60 GHz. This inductor, however, must belarge enough to simultaneously satisfy the required reactance for bothfundamental and second harmonics. Consequently, its relatively largeinductance limits the output transistor size so the PA output power(Pout) is less than 10 dBm.

There is thus a need for a power amplifier that is suitable forintegration with mm-wave 60 GHz CMOS based radios having wide bandwidthand high efficiency and generates sufficient output power.

SUMMARY OF THE INVENTION

The present invention is a novel architecture for a fully integratedswitched-mode wideband 60 GHz power amplifier. Using an appropriatesecond-harmonic termination of its output matching network, the requiredsystematic peak current of the final stage is reduced such that the PAfunctions as a class-E/F₂ switched-mode PA at saturation. Transformersof low/moderate coupling are also utilized in the preliminary stages ofthe PA to improve the overall bandwidth. In addition, the PA exploitsthe different behavior of the output impedance matching network fordifferential mode (DM) and common mode (CM) excitations.

The PA utilizes a proper second-harmonic termination in the last stageand low/moderate magnetic coupling factor transformers in theintermediate stages to reach the best product of power added efficiency(PAE) and bandwidth. The PA is also stabilized against the combinationof DM and CM oscillation modes. The invention also provides a techniqueto stabilize transformer-based mm-wave amplifiers against various modesof undesired oscillations. The power amplifier has been fabricated instandard digital 40 nm CMOS having 17.9 dBm P_(sat). The PA can beincorporated within a wide range of applications including, for example,mobile devices.

There is thus provided in accordance with the invention, a radiofrequency (RF) signal splitter, comprising a transformer having aprimary winding coupled to a differential input signal source, a firstsecondary winding configured to generate a first differential outputsignal, a second secondary winding configured to generate a seconddifferential output signal, and a cross-connect configured to couple thefirst secondary winding to the second secondary winding.

There is also provided in accordance with the invention, a radiofrequency (RF) signal splitter, comprising a transformer having aprimary winding coupled to a differential input signal source, a firstsecondary winding configured to generate a first differential outputsignal having a first polarity port and an opposite second polarityport, a second secondary winding configured to generate a seconddifferential output signal having a first polarity port and an oppositesecond polarity port, a cross-connection configured to couple the firstpolarity port of the first secondary winding to the first polarity portof the second secondary winding, and to couple the second polarity portof the first secondary winding to the second polarity port of the secondsecondary winding, wherein the cross-connection is operative to dampencombined common mode and differential mode oscillations present on thefirst and second secondary windings.

There is further provided in accordance with the invention, a class E/F₂power amplifier, comprising one or more switch transistors, a signalsplitter including a first transformer, the first transformer comprisinga primary winding coupled to a differential input signal source, a firstsecondary winding configured to generate a first differential outputsignal, a second secondary winding configured to generate a seconddifferential output signal, a cross-connect configured to couple thefirst secondary winding to the second secondary winding, an impedancematching network coupled to an output of the splitter, the impedancematching network comprising a second transformer, the second transformercomprising a primary winding having a first inductance, a secondarywinding having a second inductance, wherein the secondary inductance andits associated capacitance are configured to resonate at a fundamentalfrequency, and wherein exciting the primary winding with a common modesignal at a second harmonic frequency causes substantially no current tobe induced on the secondary winding.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is herein described, by way of example only, withreference to the accompanying drawings, wherein:

FIG. 1 is a high level schematic of an example switched-mode poweramplifier;

FIG. 2 is a graph illustrating maximum operating frequency for differentflavors of switched-mode power amplifiers;

FIG. 3 is a diagram illustrating the schematic and layout of an example60 GHz power amplifier of the present invention;

FIG. 4 is a schematic diagram illustrating an equivalent half-circuitmodel of the output matching network for differential mode (DM);

FIG. 5 is a graph illustrating matching network efficiency versusfrequency;

FIG. 6 is a high level schematic diagram illustrating an equivalenthalf-circuit model of the output matching network for common mode (CM);

FIG. 7 is a diagram illustrating the current flow in the primary andsecondary of the output matching network transformer for CM signals;

FIG. 8 is a graph of F_(c) versus CM resonant frequency;

FIG. 9 is a graph illustrating the required L_(S) and C_(S) forclass-E/F operation versus resistive load seen by the switch transistor;

FIG. 10 is a schematic diagram illustrating the half-circuit of the PApseudo-differential stage;

FIG. 11 is a schematic diagram illustrating an example cross-connecttransformer splitter of the present invention;

FIG. 12 is a half-circuit schematic diagram illustrating the transformerfor inter-stage matching;

FIG. 13 is a schematic diagram illustrating illustrates the position ofzeros and poles of the X_(in) transfer function

FIG. 14 is a graph illustrating X_(in) versus frequency for differentvalues of k_(m);

FIG. 15A is a layout diagram illustrating a top view of the splitterportion of the present invention;

FIG. 15B is a layout diagram illustrating a bottom view of the splitterportion of the present invention;

FIG. 16 is a graph illustrating measured constant maximum gain contours;

FIG. 17 is a graph illustrating measured constant PAE contours;

FIG. 18 is a block diagram illustrating an example mobile deviceincorporating the power amplifier of the present invention; and

FIG. 19 is a block diagram illustrating an example IoT nodeincorporating the oscillator/frequency generator of the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION Benefits and Constraints ofClass-E/F Power Amplifier

A high level schematic of an example class-E/F power amplifier is shownin FIG. 1. The circuit, generally referenced 10, comprises an inputmatching network 12, switch transistor M₁ (modeled as equivalent circuit16, capacitor C_(s)-C_(out), inductor L_(s) and matching network 14connected to the antenna load R_(L).

It can be shown that the drain efficiency η_(D) of zero-voltageswitching (ZVS) PA 10 can be written in terms of a set of technologydependent parameters (R_(on), C_(out)) and a set of matching network orwaveform dependent parameters (F_(C), F_(PI), F_(I)). Equation 1 belowcan be used to better understand the tradeoffs in mm-wave designs:

$\begin{matrix}{\eta_{D} = {{1 - \frac{R_{on}I_{rms}^{2}}{V_{DD}I_{DC}}} = {1 - {\left( \frac{I_{rms}}{I_{DC}} \right)^{2}{\frac{I_{DC}}{C_{S}{\omega_{0}\left( {V_{DD} - V_{sat}} \right)}} \cdot \left( \frac{V_{DD} - V_{sat}}{V_{DD}} \right)}\left( \frac{C_{S}}{C_{out}} \right)\left( {R_{on}C_{out}} \right){\omega_{0}.}}}}} & (1)\end{matrix}$

The waveform figures of merit (FoM) are defined as the following:

$\begin{matrix}{{F_{I} = \frac{I_{rms}}{I_{DC}}},} & (2) \\{{F_{C} = \frac{I_{DC}}{C_{S}{\omega_{0}\left( {V_{DD} - V_{sat}} \right)}}},} & (3) \\{F_{PI} = {\frac{I_{peak}}{I_{DC}}.}} & (4)\end{matrix}$

Where R_(on) and C_(out) are the on-state channel resistance andoff-state output capacitance of transistor M₁, respectively. Note thatR_(on)×C_(out) is invariant to changes in the width of transistor M₁.I_(DC) and I_(rms) are defined as the average and RMS values of M₁ draincurrent, and C_(S) is the PA desired shunt capacitance to satisfy theZVS criterion. V_(sat) represents the transistor's average V_(DS) in theon-state. Note that since F_(C) should not change over the ω₀=2πf₀operating frequency, C_(S) has to reduce with increasing f₀. Thus, C_(S)limits the width of the transistor at mm-wave, which leads to a dramaticincrease in R_(on) and thus V_(sat) of the switching device.Consequently, we include the effect of V_(sat) in η_(D) and F_(C)definitions in Equations 1, 2, 3 and 4 to achieve better practicalanalytic results. V_(sat) can be calculated from the following:

P _(Loss) =V _(sat) I _(DC) =R _(on) I _(rms) ² →V _(sat) =R _(on) I_(DC) F _(I) ²  (5)

By replacing I_(DC)=F_(C)C_(S)ω₀(V_(DD)−V_(sat)) in Equation 5,

$\begin{matrix}{V_{sat} = {V_{DD}\frac{F_{C}F_{I}^{2}R_{on}C_{out}\omega_{0}}{\alpha + {F_{C}F_{I}^{2}R_{on}C_{out}\omega_{0}}}}} & (6)\end{matrix}$

where α=C_(out)/C_(S) denotes how much the required C_(S) for class-E/Foperation is occupied by the self-capacitance of transistor M₁. It isalso instructive to go a step further and calculate the class-E/F PAcharacteristics based on waveform parameters and technology parametersshown in Tables 1 and 2, respectively, below.

TABLE 1 Waveform Parameters E E/F₂ E/F₃ E/F_(2,3) E/F₄ E/F_(2,4) F_(C)3.14 1.13 3.14 2.31 2.45 0.97 F_(I) 1.54 1.48 1.52 1.47 1.55 1.46 F_(PI)2.86 3.33 3.06 2.67 3.27 3.6

TABLE 2 Technology Parameters R_(in) C_(in) R_(on) C_(out) I_(out) 27kΩ/μm 1 fF/μm 850 Ω/μm 0.6 fF/μm 0.55 mA/μm

MOS devices must satisfy two conditions for proper switched-mode PAoperation. First, the transistor cut-off frequency f_(max) should be atleast three to four times higher than f₀. For example, NMOS f_(max) isabout 250 GHz in 40-nm CMOS. Therefore, the transistors should be fastenough to turn on/off rapidly at f₀=60 GHz. Second, the transistor mustbe capable of providing the required systematic peak current duringswitching while its output capacitance C_(out) remains below C_(S).Consequently, the output current I_(out) can be expressed as follows:

$\begin{matrix}{I_{out} = {{J_{out} \cdot W} = {{\overset{\_}{I_{out}} \cdot \frac{C_{out}}{\overset{\_}{C_{out}}}} \geq I_{peak}}}} & (7)\end{matrix}$

Indeed, MOS transistor current capability (I_(out)/C_(out)) isrelatively poor and puts a limit on the maximum operating frequencyf_(m) of switched-mode PAs. By using Equation 5, F_(PI) and F_(C)definitions in Equation 7, f_(m) could be derived as follows:

$\begin{matrix}{f_{m} = {\frac{\alpha}{2\pi}\frac{\overset{\_}{I_{out}}}{\overset{\_}{C_{out}}}\frac{1}{F_{C}\left( {{F_{PI}V_{DD}} - {F_{I}^{2}\overset{\_}{R_{on}I_{out}}}} \right)}}} & (8)\end{matrix}$

The f_(m) can be increased by migrating to a more advanced technology orby using a matching network with lower F_(PI) and F_(C). FIG. 2 predictsf_(m) for different flavors of class-E/F PA by utilizing Equation 8 andwaveform and technology parameters of Tables 1 and 2. FIG. 2 indicatesf_(m) can be extended to 60 GHz in 40 nm CMOS for class-E/F₂ operationwhen the effective load of the transistor is realized as an open circuitat the second harmonic 2ω₀.

By substituting Equation 6 into Equation 1, η_(D) is simplified to thefollowing:

$\begin{matrix}{\eta_{D} = \frac{\alpha}{\alpha + {F_{C}F_{I}^{2}R_{on}C_{out}\omega_{0}}}} & (9)\end{matrix}$

Equations 6 and 9 indicate that V_(sat) and η_(D) improve by using amatching network with lower F_(I) and F_(C), which is in line with f_(m)optimization. Equation 9 also predicts η_(D)=65% for a 40 nm class-E/F₂PA at 60 GHz. The switch size is relatively small such that its R_(on)degrades η_(D) to somewhere between class-A and B. It can be shown thatthe output power P_(out) of the PA can be calculated as follows:

$\begin{matrix}{P_{out} = {{\eta_{D}P_{DC}} = {\left( \frac{\alpha \; V_{DD}}{\alpha + {F_{C}F_{I}^{2}\overset{\_}{R_{on}C_{out}}\omega_{0}}} \right)^{2}F_{C}C_{S}{\omega_{0}.}}}} & (10)\end{matrix}$

The gain G_(p) of the PA can be calculated as follows:

$\begin{matrix}{G_{p} = {\frac{P_{out}}{{V_{DD}^{2}/2}\; R_{in}} = \frac{2\; \alpha \; F_{c}\overset{\_}{R_{in}C_{out}}\omega_{0}}{\left( {\alpha + {F_{C}F_{I}^{2}\overset{\_}{R_{on}C_{out}}\omega_{0}}} \right)^{2}}}} & (11)\end{matrix}$

Unfortunately, both the output power P_(out) and gain G_(p) reducealmost linearly with F_(C). Consequently, a higher f_(m) of class-E/F₂operation is achieved through painful reduction of P_(out) and preciousdevice G_(p), which can potentially reduce the total PAE.

Power Amplifier Embodiment

A high level schematic diagram illustrating an example power amplifierof the present invention is shown in FIG. 3. The circuit, generallyreferenced 20, comprises several sections including an input powersplitter 30, pre-driver amplifier stage 32, low k_(m) transformer (TRX)34, driver amplifier stage 36, cross-connect splitter of the presentinvention 38, output amplifier stage 40, series combiner 42 and parallelcombiner 44.

The PA 20 incorporates a three stage common-source pseudo-differentialpair to compensate for the gain penalty G_(p) of the class-E/F₂operation in the last PA stage 40. A transformer-based power splitter 22converts the singled-ended S_(in) input 48 to two differential signalsfeeding pre-drivers 24. Another set of splitters 28 is added before thefour parallel units of the output stage. A combination ofseries-parallel combining is used in the output matching network togenerate the output signal S_(out) 50. Two-way differential seriescombining 42 is achieved using a distributed active transformer 46 toreduce the resistive load seen by each transistor such that thesystematic P_(out) reduction of class-E/F₂ is partially compensated. Byexploiting parallel combining, the output devices can be smaller for thesame P_(out), which effectively improves the transistors' internal lossand f_(max). Hence, they can generate a stronger 2^(nd) harmoniccurrent, which is beneficial for the class-E/F₂ operation.

A high level schematic diagram illustrating an equivalent half-circuitmodel of the output matching network for differential mode (DM) is shownin FIG. 4. Circuit block 122 includes an ideal transformer (TRX) 124 andresonates at the fundamental frequency ω₀. A graph illustrating matchingnetwork efficiency versus frequency is shown in FIG. 5. The totaleffective inductance at the output of the matching network(4L_(s3)+L_(asn)+L_(asp))/2 must resonate with the parasitic capacitanceof the pad (C_(L)) to optimize its insertion loss. Furthermore, thecombination of the transformer leakage inductance L_(p3)(1−k_(m3) ²) andthe effective inductance of differential strip-lines L_(PT) along C_(S)must satisfy zero-voltage and zero-slope class-E switching criteria by:

(L _(p3)(1−k _(m3) ²)+L _(PT))C _(S)=1/4.74ω₀ ²  (12)

A high level schematic diagram illustrating an equivalent half-circuitmodel of the output matching network for common mode (CM) is shown inFIG. 6. It is important to note that the behavior of the impedancematching network is entirely different for common-mode (CM) inputsignals compared with that for DM signals. A diagram illustrating thecurrent flow in the primary 156 and secondary 154 of the output matchingnetwork transformer 152 for CM signals is shown in FIG. 7. Thetransformer coupling factor k_(m) is negligible in CM excitation andthus R_(L), C_(L) and L_(s3) cannot be seen by even harmonics. Hence,the transmission line and transformer primary inductance, which is seenby CM signals, has to resonate with C_(S) at the second harmonic 2ω₀ tosatisfy class-E/F₂ operation. The graph of F_(c) versus CM resonantfrequency of FIG. 8 indicates that F_(c) is just slightly degraded whenCM resonance lies 25% away from 2ω₀. Thus, the PA is quite insensitiveto the precise CM inductance value, which facilitates a wide bandwidthoperation.

A graph illustrating the required L_(S) and C_(S) for class-E/Foperation versus resistive load seen by the switch transistor is shownin FIG. 9. The graph shows the optimum required class-E/F₂ PA shuntcapacitance C_(S) and series inductance L_(S) at fundamental frequencyversus the load resistance presented by the matching network. Thematching network geometry design is initiated by choosing the switchtransistor dimension such that its output capacitor absorbs the entireC_(S). C_(S), however, also depends on L_(S) and the load resistancepresented by the matching network, as can be gathered from FIG. 9.Hence, several iterations are needed to find the optimal sizecombination of the transistor, transformer and matching network. Thisprocedure results in an optimal unit power transistor size of 96 (1μm/40 nm) with 1.3 dB insertion loss of the output matching network.Note that the class-E/F₂ optimal combination is different from the goalof maximizing the output power or gain.

With reference to FIG. 3, two transistor based power splitters alongwith additional series inductance and a differential strip-line arerespectively added at the input of the pre-driver 32 and driver 36stages to feed differential pairs and provide their required sourcereactance.

A schematic diagram illustrating the half-circuit of the PApseudo-differential stage is shown in FIG. 10. Each pseudo-differentialpair along with their parasitic capacitance C_(gd) and matching networkscan potentially act as two coupled Pierce oscillators and create CMinstability. It can be shown that its resonant frequency is very closeto the operating frequency (≈0.7-0.8ω₀) such that neither (1) adding anRC stabilization network at the MOS gate nor (2) matching network losscan dampen the oscillation without affecting the precious power gain atω₀. Fortunately, using relatively large resistors (R_(B)˜3 kΩ) betweenthe center tap of the secondary windings of the input and inter-stagetransformers and gate bias voltage can cancel out the CM currents at thetransformer secondary winding. Hence, any CM oscillation will bedampened. Nevertheless, a combination of CM and differential mode (DM)oscillation can potentially happen in the transformer splitter, i.e.cross-connect splitter 28 (FIG. 1).

A schematic diagram illustrating an example cross-connect transformersplitter of the present invention is shown in FIG. 11. The well-knownneutralization technique can be used to improve DM stability and gainsensitivity to the load impedance. The PA is stabilized for CMoscillation by use of relatively large resistors (e.g., R_(B)˜3 kΩ)between the center tap of the secondary windings of the transformers andgate bias voltage. Thus, the CM currents are canceled out at thetransformer secondary and CM oscillation is dampened. A parallel RCnetwork (e.g., 51Ω∥144 fF) is also added at the input of the first stageto introduce resistive loss at lower frequencies. Nevertheless, acombination of CM and DM oscillation can potentially occur in thetransformer splitter. As shown, each differential pair 181, 183 coupledto the transformer splitter 182 could oscillate in CM but with 180°phase shift to each other. Hence, neither neutralization capacitors norR_(B) will damp this oscillation. The present invention provides asolution by adding a relatively weak cross connection 188 between thesplitter's in-phase ports to reduce the loop gain of this oscillationmode without affecting the splitter's main function. In anothersolution, a lossy path is added between the ground connections of twopseudo-differential pairs across the splitter. In one embodiment, thewidth of the added cross connection is preferably at least four timesnarrower than the main metal line of the splitter. It is appreciated,however, that the invention it not limited to the cross connectiondisclosed herein and can be modified depending on the particularimplementation.

The effective Q-factor of the PA input/output matching network isdegraded by the 50Ω load and RF pad parasitic capacitance, C_(L)≦50 fF,to about 1 to 2 at 60 GHz, thus making these networks wideband. Theinput impedance of MOS transistors, however, is considered as load tothe inter-stage matching network, where Qeff= R_(in)C_(in) ω₀≈60 GHz.Hence, the impedance seen at the input of the transformer network(r_(in)+jX_(in) in FIG. 12) changes significantly over frequency andthus limits the bandwidth of the PA. FIG. 13 illustrates the position ofzeros and poles of the X_(in) transfer function. In the case of a highk_(m) (≧0.7), the conjugate zeros pair occurs at much higher frequencythan the poles of the system. Hence, a large variation is seen inX_(in), as shown in FIG. 14 where trace 200 represents X_(in) fork_(m)=0.75; trace 202 represents X_(in) for k_(m)=0.5; trace 204represents X_(in) for k_(m)=0.25. The zero/pole pairs, however, comecloser together with lower k_(m) resulting in a flatter region beingobserved in the X_(in) plot. Hence, the transistor sees its desiredimpedance over a wider frequency range. The additional insertion losspenalty is only ≦1.5 dB over the bandwidth by using a k_(m)=0.25transformer. That penalty happens at the primary stages where it has anegligible effect on the total PAE.

Performance Results

The mm-wave power amplifier of the present invention has been fabricatedin an integrated circuit using 40 nm 1.1V CMOS technology. A layoutdiagram illustrating a top view of the splitter portion of the presentinvention is shown in FIG. 15A. A layout diagram illustrating a bottomview of the splitter portion of the present invention is shown in FIG.15B. The circuit, generally referenced 260, comprises the primarywinding 262 of the splitter, first secondary winding 266, secondsecondary winding 264, a connection 270 connecting the positive polarityports of the secondary windings 264, 266, a connection 268 connectingthe negative polarity ports of the secondary windings 264, 266 andV_(DD) trace 274.

In one embodiment, the transformers comprising the PA are completelyfilled with dummy metal strips to comply with the strict metal densityrules. The amount of the metal fills right underneath the transformerwindings is kept at minimum to reduce the extra parasitic capacitanceand eddy current losses. Electromagnetic simulations, however, reveal anadditional loss of 0.2 to 0.4 dB for each matching network.

With a 1 V supply, the PA achieves a peak power gain of 21.6 dB at 58GHz with a 3 dB bandwidth of 9.7 GHz (i.e. 51.5 to 61.2 GHz). The S₁₁,S₂₂ and S₁₂ are respectively better than −6, −7 and −42 dB within 50 to67 GHz. The large-signal measurements were performed by a mixed-signalactive load-pull setup. Consuming ≦0.3 A from a 1 V supply, the measuredP_(1dB) and P_(sat) are respectively 14.9 dBm and 17.9 dBm with 20.5%PAE. at 60 GHz. The following parameters are maintained over 52 to 63GHz: 16.9 dBm P_(sat), 13.8 dBm P_(1dB), and 16% PAE. FIGS. 16 and 17illustrate the constant gain and PAE contours of the power amplifier,respectively, and also verifies its stability over load variation.

Mobile Device Incorporating the Power Amplifier

A block diagram illustrating an example tablet/mobile deviceincorporating the power amplifier of the present invention is shown inFIG. 18. The mobile device is preferably a two-way communication devicehaving voice and/or data communication capabilities. In addition, thedevice optionally has the capability to communicate with other computersystems via the Internet. Note that the mobile device may comprise anysuitable wired or wireless device such as multimedia player, mobilecommunication device, digital still or video camera, cellular phone,smartphone, iPhone, PDA, PNA, Bluetooth device, tablet computing devicesuch as the iPad or other iOS device, Android device, Surface, Nexus,Google Glass, etc. For illustration purposes only, the device is shownas a mobile device, such as a cellular based telephone, smartphone orsuperphone. Note that this example is not intended to limit the scope ofthe mechanism as the invention can be implemented in a wide variety ofcommunication devices. It is further appreciated the mobile device shownis intentionally simplified to illustrate only certain components, asthe mobile device may comprise other components and subsystems beyondthose shown.

The mobile device, generally referenced 370, comprises one or moreprocessors 400 which may comprise a baseband processor, CPU,microprocessor, DSP, etc., optionally having both analog and digitalportions. The mobile device may comprise a plurality of cellular radios430 and associated antennas 432. Radios for the basic cellular link andany number of other wireless standards and Radio Access Technologies(RATs) may be included. Examples include, but are not limited to, ThirdGeneration (3G) Long Term Evolution (LTE), Code Division Multiple Access(CDMA), Personal Communication Services (PCS), Global System for MobileCommunication (GSM)/GPRS/EDGE 3G; WCDMA; WiMAX for providing WiMAXwireless connectivity when within the range of a WiMAX wireless network;Bluetooth for providing Bluetooth wireless connectivity when within therange of a Bluetooth wireless network; WLAN for providing wirelessconnectivity when in a hot spot or within the range of an ad hoc,infrastructure or mesh based wireless LAN (WLAN) network; near fieldcommunications; UWB; GPS receiver for receiving GPS radio signalstransmitted from one or more orbiting GPS satellites, FM transceiverprovides the user the ability to listen to FM broadcasts as well as theability to transmit audio over an unused FM station at low power, suchas for playback over a car or home stereo system having an FM receiver,digital broadcast television, etc.

The mobile device may also comprise internal volatile storage 436 (e.g.,RAM) and persistent storage 440 (e.g., ROM) and flash memory 438.Persistent storage 436 also stores applications executable byprocessor(s) 400 including the related data files used by thoseapplications to allow device 370 to perform its intended functions.Several optional user-interface devices include trackball/thumbwheelwhich may comprise a depressible thumbwheel/trackball that is used fornavigation, selection of menu choices and confirmation of action,keypad/keyboard such as arranged in QWERTY fashion for enteringalphanumeric data and a numeric keypad for entering dialing digits andfor other controls and inputs (the keyboard may also contain symbol,function and command keys such as a phone send/end key, a menu key andan escape key), headset 388, earpiece 386 and/or speaker 384,microphone(s) and associated audio codec 390 or other multimedia codecs,vibrator for alerting a user, one or more cameras and related circuitry420, 422, display(s) 434 and associated display controller 426 andtouchscreen control 424. Serial ports include a micro USB port 378 andrelated USB PHY 376 and micro SD port 380. Other interface connectionsmay include SPI, SDIO, PCI, USB, etc. for providing a serial link to auser's PC or other device. SIM/RUIM card 382 provides the interface to auser's SIM or RUIM card for storing user data such as address bookentries, user identification, etc.

Portable power is provided by the battery 374 coupled to powermanagement circuitry 372. External power is provided via USB power or anAC/DC adapter connected to the power management circuitry that isoperative to manage the charging and discharging of the battery. Inaddition to a battery and AC/DC external power source, additionaloptional power sources each with its own power limitations, include: aspeaker phone, DC/DC power source, and any bus powered power source(e.g., USB device in bus powered mode).

Operating system software executed by the processor 400 is preferablystored in persistent storage (i.e. ROM 440), or flash memory 438, butmay be stored in other types of memory devices. In addition, systemsoftware, specific device applications, or parts thereof, may betemporarily loaded into volatile storage 436, such as random accessmemory (RAM). Communications signals received by the mobile device mayalso be stored in the RAM.

The processor 400, in addition to its operating system functions,enables execution of software applications on the device 370. Apredetermined set of applications that control basic device operations,such as data and voice communications, may be installed duringmanufacture. Additional applications (or apps) may be downloaded fromthe Internet and installed in memory for execution on the processor.Alternatively, software may be downloaded via any other suitableprotocol, such as SDIO, USB, network server, etc.

Other components of the mobile device include an accelerometer 418 fordetecting motion and orientation of the device, gyroscope 417 formeasuring or maintaining orientation, magnetometer 416 for detecting theearth's magnetic field, FM radio 412 and antenna 413, Bluetooth radio408 and antenna 410, Wi-Fi radio 398 including antenna 402 and GPS 392and antenna 394.

In accordance with the invention, the mobile device 370 comprises one ormore PA circuits, each incorporating the power amplifier circuit of thepresent invention described in detail supra. Numerous embodiments of themobile device 370 may comprise a PA circuit 428 as described supraincorporated in the one or more cellular radios 430; a PA circuit 414 asdescribed supra incorporated in the FM radio 412; a PA circuit 406 asdescribed supra incorporated in the Bluetooth radio 408; a PA circuit404 as described supra incorporated in the Wi-Fi radio 398; and a PAcircuit 396 as described supra incorporated in the GPS radio 392.

Internet of Things (IoT) Node Incorporating the Power Amplifier

The Internet of Things (IoT) is defined as the network of physicalobjects or “things” embedded with electronics, software, sensors andnetwork connectivity, which enables these objects to collect andexchange data. The IoT allows objects to be sensed and controlledremotely across existing network infrastructure, creating opportunitiesfor more direct integration between the physical world andcomputer-based systems, and resulting in improved efficiency, accuracyand economic benefit. Each thing is uniquely identifiable through itsembedded computing system but is able to interoperate within theexisting Internet infrastructure. Experts estimate that the IoT willconsist of almost 50 billion objects by 2020.

A block diagram illustrating an example IoT node incorporating theoscillator/frequency generator of the present invention is shown in FIG.19. The example IoT, generally referenced 950, comprises a plurality ofnodes 990. The architecture of an example IoT node 952 shown can befully integrated as a System on Chip (SoC) on a single IC chip innanoscale CMOS. It contains the radio subsystem to wirelesslycommunicate with other nodes and gateways 992, application processor toimpart a certain amount of local “intelligence”, sensor and an optionalactuator to interface with the environment and energy management toharvest energy (light, heat, vibration or RF power) from the environmentand/or convert the voltage levels to those required by the functionalcircuitry. The RF and non-RF frequency synthesizers provide localoscillator and processor clocks, respectively. A frequency reference 994provides a fixed clock with excellent long term stability to thefrequency synthesizers. In one embodiment, the power amplifier of thepresent invention described supra is incorporated in the RF transceiver958 as circuit block 970.

The RF transceiver 958 interfaces with an antenna 956. The RF signalsare upconverted and downconverted there to the lower (i.e. baseband)frequencies, which are then processed in the analog baseband circuitry.The conversion from analog to digital (i.e. ADC), and vice versa (i.e.DAC), is also performed there. The digital baseband completes thephysical layer of a chosen communication standard. The applicationprocessor performs various control and signal processing functions andis responsible for giving a level of “intelligence” to the IoT node.

The RF frequency synthesizer 954 is realized as an all-digital PLL(ADPLL) and provides a local oscillator signal to the RF transceiver958. The non-RF frequency synthesizer 964 provides clocks to the digitalbaseband 962 and application processors 974. The clock frequency has tobe dynamically switchable in response to the changing computational loadconditions. The energy management (EM) circuitry 972 provides energyconversion between the energy harvester 978 and/or low-capacity storagebattery 980 and all the IoT functional circuits. The EM circuit carriesout several functions. First, it boosts the voltage from the energyharvester (e.g., light, heat, vibration, RF electromagnetic, etc.) tothat required by the nanoscale CMOS circuits, which is in the range of0.7 to 1.0 V assuming 40 nm CMOS technology. This is performed by adedicated DC-DC boost converter 976. Second, it down-shifts the energyfrom a battery, which is on the order of 1.5 to 3.6 V to that requiredby the nanoscale CMOS circuits. This is performed by a dedicated DC-DCbuck converter 976. Third, both boost and buck converters use energystorage passive devices, i.e. capacitor or inductor for storingelectrical and magnetic energy, respectively, in order to change thevoltage level with high efficiency. The high conversion efficiency mustbe maintained across the entire range of the allowed loads. Fourth, theEM needs to provide many power supply domains. This is dictated by thedifferent voltage level requirements during voltage scaling. Fifth, theEM supply domains preferably provide individually adjustable voltagelevels. The supply voltage level of digital logic circuits widely varydepending on the fast changing real time computational load conditions,while the voltage level of digital RF and analog circuits experienceless of such variance, and mainly due to temperature and operatingfrequency, as well as communication channel conditions. Moreover, theanalog circuits have to be properly biased, which normally prevents themfrom operating at near-threshold conditions.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the invention. Asused herein, the singular forms “a”, “an” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises”and/or “comprising,” when used in this specification, specify thepresence of stated features, integers, steps, operations, elements,and/or components, but do not preclude the presence or addition of oneor more other features, integers, steps, operations, elements,components, and/or groups thereof.

The corresponding structures, materials, acts, and equivalents of allmeans or step plus function elements in the claims below are intended toinclude any structure, material, or act for performing the function incombination with other claimed elements as specifically claimed. Thedescription of the present invention has been presented for purposes ofillustration and description, but is not intended to be exhaustive orlimited to the invention in the form disclosed. As numerousmodifications and changes will readily occur to those skilled in theart, it is intended that the invention not be limited to the limitednumber of embodiments described herein. Accordingly, it will beappreciated that all suitable variations, modifications and equivalentsmay be resorted to, falling within the spirit and scope of the presentinvention. The embodiments were chosen and described in order to bestexplain the principles of the invention and the practical application,and to enable others of ordinary skill in the art to understand theinvention for various embodiments with various modifications as aresuited to the particular use contemplated.

What is claimed is:
 1. A radio frequency (RF) signal splitter,comprising: a transformer having: a primary winding coupled to adifferential input signal source; a first secondary winding configuredto generate a first differential output signal; a second secondarywinding configured to generate a second differential output signal; anda cross-connect configured to couple said first secondary winding tosaid second secondary winding.
 2. The signal splitter according to claim1, wherein said cross connection is placed between in-phase ports ofsaid first and second secondary windings.
 3. The signal splitteraccording to claim 1, wherein said cross connection is operative toreduce loop gain of a combination of differential mode and common modeoscillations.
 4. The signal splitter according to claim 1, wherein saidcross connection does not affect normal operation of said splitter. 5.The signal splitter according to claim 1, wherein said cross connectionis operative to introduce signal loss in a return of said first andsecond differential output signals.
 6. A radio frequency (RF) signalsplitter, comprising: a transformer having: a primary winding coupled toa differential input signal source; a first secondary winding configuredto generate a first differential output signal having a first polarityport and an opposite second polarity port; a second secondary windingconfigured to generate a second differential output signal having afirst polarity port and an opposite second polarity port; across-connection configured to couple the first polarity port of saidfirst secondary winding to the first polarity port of said secondsecondary winding, and to couple the second polarity port of said firstsecondary winding to the second polarity port of said second secondarywinding; wherein said cross-connection is operative to dampen combinedcommon mode and differential mode oscillations present on said first andsecond secondary windings.
 7. The signal splitter according to claim 6,wherein said cross connection comprises a relatively weak electricalconnection.
 8. The signal splitter according to claim 6, wherein saidcross connection is placed between in-phase ports of said first andsecond secondary windings.
 9. The signal splitter according to claim 6,wherein said cross connection is operative to reduce loop gain of acombination of differential mode and common mode oscillations.
 10. Thesignal splitter according to claim 6, wherein said cross connection doesnot affect normal operation of said splitter.
 11. The signal splitteraccording to claim 6, wherein said cross connection is operative tointroduce signal loss in a return of said first and second differentialoutput signals.
 12. A class E/F₂ power amplifier, comprising: one ormore switch transistors; a signal splitter including a firsttransformer, said first transformer comprising: a primary windingcoupled to a differential input signal source; a first secondary windingconfigured to generate a first differential output signal; a secondsecondary winding configured to generate a second differential outputsignal; a cross-connect configured to couple said first secondarywinding to said second secondary winding; an impedance matching networkcoupled to an output of said splitter, said impedance matching networkcomprising: a second transformer, said second transformer comprising: aprimary winding having a first inductance; a secondary winding having asecond inductance; wherein said secondary inductance and its associatedcapacitance are configured to resonate at a fundamental frequency; andwherein exciting said primary winding with a common mode signal at asecond harmonic frequency causes substantially no current to be inducedon said secondary winding.
 13. The power amplifier according to claim12, wherein said cross connection is placed between in-phase ports ofsaid first and second secondary windings.
 14. The power amplifieraccording to claim 12, wherein said cross connection is operative toreduce loop gain of a combination of differential mode and common modeoscillations.
 15. The power amplifier according to claim 12, whereinsaid cross connection does not affect normal operation of said splitter.16. The power amplifier according to claim 12, wherein said crossconnection is operative to introduce signal loss in a return of saidfirst and second differential output signals.